Linear FM radar

ABSTRACT

A FM-CW radar system comprises a frequency modulated continuous wave digital generator that produces both in-phase (I) and quadrature-phase (Q) outputs to orthogonally oriented transmitter antennas. A linearly polarized beam is output from a switched antenna array that allows a variety of I-and-Q pairs of bowtie antennas to be alternately connected to the transmitter and receiver. The receiver inputs I-and-Q signals from another bowtie antenna in the array and mixes these with samples from the transmitter. Such synchronous detection produces I-and-Q beat frequency products that are sampled by dual analog-to-digital converters (ADC&#39;s). The digital samples receive four kinds of compensation, including frequency-and-phase, wiring delay, and fast Fourier transform (FFT). The compensated samples are then digitally converted by an FFT-unit into time-domain signals. Such can then be processed conventionally for range information to the target that has returned the FM-CW echo signal.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to radar and ground penetrating radarimaging methods, and more particularly to radars that linearly sweepthrough frequencies and synchronously detect return signals affected bytime-of-flight delays to target objects.

2. Description of Related Art

Frequency modulated (FM) continuous wave (CW) radars transmit a sweepfrequency that is mixed with a return echo to produce a beat-frequency.The beat frequency output from the mixer is a function of both how fastthe CW output is sweeping in frequency and how far the return echo hadto travel from the transmitter to the target and back to the receiver. Afaster FM sweep of the CW signal increases the scale of the beatfrequency product. Given a linear sweep, e.g., a sawtooth, and a fixeddistance to the radar target, then the beat frequency will be a steadytone. The bandwidth of that tone determines the resolution of the radar.

The beat frequency tone represents the range of the target in thefrequency domain. Traditional radars launch radio frequency (RF) pulsesthat are delayed in their echo returns in time by how far they had tofly out and back. So traditional radars produce range signals in thetime domain. A fast Fourier transform (FFT) is typically used to convertFM-CW radar frequency-domain range signals to time-domain range signalsso they can be conventionally processed.

Yukinori Yamada describes an FM-CW radar in U.S. Pat. No. 6,121,917,issued Sep. 19, 2000. An array antenna, beat signals, and Fouriertransform process are used to Fourier transform data from each beamangle. Such radar determines the range to an object near the antennaarray.

Yukinori Yamada describes another FM-CW radar apparatus in U.S. Pat. No.6,445,339 B1, issued Sep. 3, 2002. The transmitted signal used is afrequency modulated continuous wave. A beat signal is generated frommixing transmitted and received signals, and this is the A/D converted.A switch is used to select various antenna elements in an array. Adigital signal processor executes a digital beam-forming operation todetect the target from the beat signals.

The phase delays imparted by electronic FFT devices are not constant andvary with frequency. In order to get a accurate conversion between thefrequency and time domains, the FFT output needs to be appropriatelyphase compensated. But to do this, the frequency of the signal beingprocessed must be known to apply the appropriate correction. InFM-radar, the frequency of the return echo signal is unpredictablebecause it depends on the unknown range to the target. Prior art hasneither recognized this source of error nor have there been anysolutions proposed in conventional radar implementations.

SUMMARY OF THE INVENTION

Briefly, a FM-CW radar system embodiment of the present inventioncomprises a frequency modulated continuous wave digital signal generatorthat produces both in-phase (I) and quadrature-phase (Q) outputs, twopairs of bowtie transmit and receive antennas orthogonally placed on arotating platform, an antenna switch matrix for routing the transmitsignal from the digital signal generator to the desired transmit antennaand for routing the output of the desired receive antenna to the radarreceiver input. A dual channel radar receiver is provided which mixesthe received signal synchronously with the I and Q outputs from thedigital signal generator. Such synchronous detection produces I-and-Qbeat frequency products that are sampled by dual analog-to-digitalconverters (ADC's). These digital samples receive four kinds ofcompensation, including dynamic frequency-and-phase, static wiringdelay, and novel fast Fourier transform (FFT) filter phase corrections.The digital samples are then converted by an FFT-unit into precisecoherent time-domain signals. Such coherent time domain signals taken atfine sample intervals over the surface of the ground can then beprocessed by conventional back projection techniques to yield3-dimensional images of the underground structures that returned theFM-CW echo signal.

By a combination of the linear motion of the radar antenna platformrotation axis along the ground and the rotary motion of the antennasabout this axis a very fine sampling of radar echo data in the X-Y planeis obtained at sample spacings much less than the antenna size can berapidly obtained over a considerable swath width for both orthogonalpolarizations.

An advantage of the present invention is that a digital linear-FMground-penetrating radar is provided that is less bulky, easier tomaneuver, and provides finer radar details compared to multi-antennafixed arrays.

Another advantage of the present invention is that a means is providedthat can collect radar data at intervals smaller than the size of theantenna both along and perpendicular to the direction of motion of theradar along the ground because the radar antenna is also moved in acircle.

A further advantage of the present invention is that a digital linear-FMground-penetrating radar is provided that can provide sharp 3-Dsubsurface images over a substantial swath width in a single pass of theradar.

A still further advantage of the present invention is that the digitallinear-FM ground-penetrating radar is produces higher spatial resolutionimages due to the set of phase and amplitude compensations that areapplied.

Another advantage of the present invention is that a digital linear-FMground-penetrating radar is provided that is more efficient and haslower power consumption due to digitizing the radar data in thefrequency domain rather than the time domain.

The above and still further objects, features, and advantages of thepresent invention will become apparent upon consideration of thefollowing detailed description of specific embodiments thereof,especially when taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram of an FM-CW radar system embodimentof the present invention shown transmitting a signal that is reflectedback by a radar target;

FIG. 2A is a bottom-view diagram of a rotating antenna array disc withswitchable transmitting and receiving bowtie antenna pairs that arerotated in a plane, e.g., parallel to the ground surface in anearth-penetrating radar application;

FIG. 2B is a side-view diagram of the rotating antenna array disc ofFIG. 2A, as it is mounted inside a radar-absorbing shroud and rotated byan axle motor;

FIG. 3 represents the different phase and amplitude responses of FFTfilters at various frequencies measured by the FM-CW radar system ofFIG. 1;

FIG. 4 is a functional block diagram of a first method for correctingFFT phase errors that uses two different size FFT's with different phaseslopes; and

FIG. 5 is a functional block diagram of a second method for correctingFFT phase errors that compares the amplitude responses of two adjacentFFT filters.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates an FM-CW radar system embodiment of the presentinvention, and is referred to herein by the general reference numeral100. The radar system 100 comprises a frequency modulator (FM) 102 thatcauses a continuous wave (CW) generator 104 to linearly sweep through aband of frequencies. For example, at a time-1 (t1) the transmitterfrequency from CW generator 104 will be frequency-1 (f1). At a time-2(t2), the transmitter frequency will slew up to a frequency-2 (f2). Andat a time-3 (t3), the transmitter frequency will slew further to afrequency-3 (f3). An in-phase (I) unit 106 digitally produces anI-signal, and a quadrature-phase (Q) unit 108 digitally produces aQ-signal 90-degrees shifted in phase. The I-signal is amplified by apower amplifier 110 before being selectively switched through an antennamatrix 112 to a rotating antenna array 114.

The rotating antenna array 114 allows, e.g., transmitting antennas 116and 118, to radiate their signals from many finely separated positionsin a plane of space. Radar targets of interest are approached normal tothe plane of rotation. Radar data collected from each spatial positionvisited by the antennas allows for very high resolutionthree-dimensional radar images to be topographically computed. Thethree-dimensional spatial locations of each antenna at the times thesignals are transmitted and received are reported and/or computed usingconventional techniques, e.g., dead-reckoning, shaft encoders, LIDAR,GPS, etc. These positions are associated and correlated with the radarsignals for tomography.

The transmitting antennas 116 and 118 produce a linearly polarizedtransmission signal 120 that is directed toward a radar target 122. FIG.1 arbitrarily assumes that the time-of-flight propagation delay oftransmission signal 114 will delay its arrival at radar target 116 untiltime t2. By time t2, the transmitter signal being output by FM 102 andCW 104 will have swept to frequency f2. And, by time t3, the transmittersignal being output by FM 102 and CW 104 will have swept to frequencyf3.

The return flight of an echo signal 124 will experience a similarpropagation time delay. So it will not arrive as frequency f1 until atime t3 at a pair of receiving antennas 126 and 128. After beingselected by matrix 112, the received signals are amplified by alow-noise amplifier (LNA) 130. A Q-sampler 131 provides a quadraturelocal oscillator (LO) demodulation reference for a Q-mixer 132. AnI-sampler 133 provides an in-phase local oscillator (LO) demodulationreference for an I-mixer 134. These analog signals are digitally sampledby analog-to-digital converters (ADC) 136 and 138.

Each of the mixers 132 and 134 produces a beat tone that represents thedifference in frequencies between the outgoing signal 120 and the returnecho 124. The frequency of the beat tone depends on the distance toradar object 122. Minor cable, device, and wiring delays are ignoredhere, but in practice are compensated or nulled out.

Since it is the beat products from the mixers 132 and 134 that representuseful data, the range of frequencies are quite modest that must beconverted by ADC's 136 and 138. In practical implementations, 14-bitbinary ADC's are very affordable and perform very well. The digitaloutputs are connected to provide data to a digital signal processor(DSP) 140.

Software within the DSP 140 is used to implement a compensation function142, fast Fourier transformation 144, and FFT corrections process 146.As a result, a time-domain radar output 148 provides range data for theradar object 116. Basically, the higher the beat tone, the farther awayis the radar object 116. This has to be converted into a complex timepulse with a phase that is proportional to the distance to the radarobject 122.

FIG. 2A represents a radar system 200 with a rotating antenna array disc202 nested within a radar-absorbing shroud 204. A motor turns the disc202 within the shroud 204, the direction of rotation is unimportant.Switchable transmitting and receiving bowtie antenna pairs 206 and 208are rotated on an axis 210 in a plane, e.g., parallel to the groundsurface in an earth-penetrating radar application. The bowtie antennapairs 206 and 208 correspond to antennas 116, 118, 126, and 128, inFIG. 1. The whole assembly can be carried along a path while theantennas are rotating. An ideal combination of path speed, rotationalspeed, and frequency of radar transmissions can result in manymeasurement points individually separated by less than an inch in space.

In a prototype embodiment that was built, the antenna array disc 202 wasforty inches (40″) in diameter. Each antenna pair 206 and 208 was in afield 15″ square, with each bowtie element being about 5¾″ by 12″.Matching bowtie elements were separated by 8″. A radar box and computerbox were mounted to ride along with the antennas on the disc. Suchsimplified wiring. These used WiFi transmissions to communicatemeasurements over a wireless network to a local computer. The imageswere rendered on the computer. Operating power was supplied through sliprings.

FIG. 2B shows radar system 200 from the side with antenna array disc 202rotating horizontally and normal to the page. The radar-absorbing shroud204 protects the antennas from emitting or receiving spurious signalsfrom the sides or top. A motor 212 turns the disc 202 within the shroud204, and can be attached to stationary objects to scan moving targets,or moving objects to scan stationary and moving targets, e.g., a tripod,a wall, a gateway, a roadway, a boom arm, an aircraft, a vehicle, acrane, etc. An encoder 214 reports the shaft angle of axis 210.

A radar unit 216 rides along on disc 202 with antennas 206 and 208. Itwirelessly communicates its measurements to a WiFi receiver 218. Forexample, a pair of radar targets 220 and 222 echo signals back, andtheir relative locations are measured by radar unit 216. Over time, manysuch measurements can be collected as the disc rotates and thegeometries change to allow different perspectives. The otherwiseone-dimension measurements of the radar echo returns can then be used topaint a high-resolution three-dimensional picture as the antennapositions are correlated to the measurements obtained.

Referring again to FIG. 1, the FFT corrections process 146 runs withinDSP 140. The various frequency and phase errors that are caused by thephysical implementations of the transmitter, antennas, and receiver arenulled out by the compensation function 142. But, the FFT 144 injectsphase errors that are a function of the input frequency, e.g., the beattone. Since the beat tone is of an unknown frequency and dynamic, itcannot be simply indexed to find an appropriate FFT phase errorcorrection factor.

It has been observed in experiments that the FFT phase errors aresubstantial and the time-domain radar output 148 will be grosslyinaccurate if such errors are not corrected. The FM-CW radar system 100effectively produces a steady beat tone as the FM sweep progresses overthe sweep bandwidth if the antenna-target geometry is unchanging. Allthe samples taken by the ADC's 136 and 138 can be effectively averagedto arrive at a high confidence range estimate. But these samples willnot vector-add in a single consistent direction if each sample is lateradversely affected by differing FFT phase-error injections in FFT 144.

FIG. 3 represents the phase and amplitude responses of two FFT filters,A and B, by frequency. The phase and amplitude responses of the FFTfilters is different at each frequency point.

FIG. 4 represents a first method for compensating the phase errorscaused by the FFT. A subsystem 400 inputs the detected differencefrequencies 402 from the mixer into an anti-aliasing filter 404. Suchfilter is essentially a low-pass filter with a very sharp cut-offfrequency. An analog-to-digital converter converts these signals intoanalog form.

In one embodiment of the present invention that was built as a prototypeand demonstration unit, a digital linear-FM core technology module (CTM)was implemented as the central element in at least two commercialground-penetrating radar products, e.g., a handheld field locator, and avehicle-mounted road survey unit. The CTM uses direct digital synthesisradio-frequency (RF) technology to generate linear frequency modulated(linear-FM) radar waveforms. It integrated all the functions necessaryto interface between a digital computer and the analog signals of theradar transmitter and receiver. The CTM included a direct digitalsynthesizer (DDS) for generating in-phase and quadrature linear-FMwaveforms. Twin programmable anti-aliasing filters and 14-bitanalog-to-digital converters convert the output of the radar receiverinto digital form which is directly stored in the memory of a powerfuldigital signal processor (DSP). The CTM interfaces the DSP chip to adigital computer via a high-speed parallel bus. Such allows downloadingof the DSP operating program and operating commands, and uploading ofany processed radar data.

The DSP signal processing transforms frequency-domain linear-FM videodata into a series of time-domain echo samples ready for 3-D imageformation. The DSP accepts raw radar data from the analog-to-digitalconverters, subtracts any internal clutter, applies phase and amplitudeweighting to the video data, and applies a Fourier transform to producethe output data samples.

The CTM includes power management circuitry to minimize powerconsumption operation in remote locations using battery power. The CTMis essentially digital and therefore easily manufacturable usingconventional surface mount manufacturing techniques. Testing can then bedone without special test equipment, e.g., the CTM has no potentiometersor other components requiring manual adjustment.

Such digital linear-FM (DLFM) radar technology has major advantages overswept frequency and the commercially more common impulse-based radars.For example, an improved ability to see small objects in the presence oflarge objects, improved detection of targets/objects from which a lowsignal is received and must compete with internal radar-receiver-noise,sharper images with fewer signal processing steps, lower powerconsumption, programmable to deliver specific image and map data needs,greater reliability of operation, compliance with existing FCC Part 15rules for unlicensed operation, more thorough rejection of internalclutter signals, reduced interference to and from other systems sharingthe spectrum, and better resistance to image distortion and positioningerrors that result from antenna ringing effects.

A major advantage of DLFM radar technology is that it allows precisecontrol over the shape of the radiated spectrum as a function offrequency. This precise control of spectrum shape translates into theimproved detection of low signal-to-noise underground objects and theinherent ability to conform to present FCC Part 15 regulations.

The improved detection of low signal-to-noise ratio underground objectsresults because the linear-FM waveform can arbitrarily shape itstransmitted signal strength as a function of frequency. In the simplestcase, a linear-FM system transmits the same power at each frequencywithin its operating band. The flat spectrum of the transmitted signalmatches the flat spectrum of receiver noise, and so the signal-to-noiseof each spectral component is the same.

In impulse radar, the spectrum of the transmitted signal is not flat,and the signal-to-noise ratio of each spectral component varies withfrequency, having a maximum at some center frequency and falling oneither side of this frequency. This means that given the sametransmitted energy the effective signal-to-noise ratio of a linear-FMradar will be greater than that of an impulse radar. This also meansthat the effective bandwidth of the received signal depends on itssignal-to-noise ratio. At low signal-to-noise ratios, the radar rangeresolution of an impulse radar degrades relative to that of a linear-FMradar.

The spectrum of an impulse radar is determined by the Fourier transformof the shape of the transmitted impulse, so impulse radars radiatesignificant energy over a very wide bandwidth. Such energy decreasesvery little in strength from the center frequency. There is no sharpcutoff in the spectrum of radiated energy. The only way to limit theradiated energy above an arbitrary frequency, such as mandated by FCCregulations, is to either lower the impulse center frequency or to passthe impulse signal through a sharp cutoff low-pass filter. The firstreduces the range resolution, and the second degrades the rangeresolution and distorts the transmitted waveform.

A linear-FM radar intrinsically has a very sharp cutoff in radiatedpower outside its operating band, and full power operation at allfrequencies within this band. It is thus simultaneously capable ofoperating very close to a specific cutoff frequency with minimumradiation in a restricted region above this frequency, and produces abetter range resolution than an impulse radar operating underrestrictions.

The radar signals radiated by a linear-FM radar are narrowband waveformswhich linearly chirp over a given frequency range rather than radiatingall frequencies at the same instant of time. This allows a reduction inthe required analog-to-digital conversion rate by a factor ofapproximately one thousand, relative to that required by an impulseradar. Such translates into lower hardware costs, lower system powerconsumption, an improved ability to see weak echoes in the presence ofstrong ones, and sharper images with fewer required signal processingcomputations.

In a linear-FM system, the received radar echo signals are mixed with asample of the transmitted signal for synchronous detection. The mixeroutput is filtered so that only the difference-frequency signals remain.A radar scatterer at a fixed range will be seen as a relatively longduration fixed-frequency sinewave signal, and the difference frequencyis proportional to radar range. The minimum sample rates needed arereduced by the range window width divided by the sweep duration and thevelocity of propagation. For a linear-FM radar with a 50-ft rangewindow, and a sweep duration of 500-microseconds, such minimum isreduced about 3,920:1.

An impulse radar with an upper frequency cutoff of 960 MHz requiresanalog-to-digital conversion rate greater than 1,920 MHz. A linear-FMradar operating with the same range resolution requires ananalog-to-digital conversion rate of 0.49 MHz. The impulse radardigitizes at an extremely high rate for a very short time, e.g.,100-nanoseconds. The linear-FM radar digitizes at a much lower rate andover a relatively longer time. A low repetition rate avoids interferencefrom multiple-time-around echoes. So linear-FM radar uses this time tosignificantly reduce the required analog-to-digital conversion rate.

Reducing the analog-to-digital rate simplifies the radar hardware sinceit no longer has to be good enough to input data samples at a gigahertzrate. Reduced performance demands lower the cost and power consumptionof the radar/computer interface hardware. The lower analog-to-digitalrate also means that analog-to-digital conversion can be done with muchgreater precision. Such converters able to operate at gigahertz ratestypically have only 8-bit precision. A 14-bit converter can be used alinear-FM system, and has about 36-dB more dynamic range.

Referring now to FIG. 4, the FFT contributed phase error can becorrected even when the input frequency is not known before hand. Thephase error responses of FFT filters with different numbers of pointsbut connected to the same input signals can be compared to discover theinput frequency they have both processed. FIG. 3 shows the two differentphase responses, A and B, of different multi-point FFT filters. At anyone frequency, the difference between the two phase slopes will beunique. Such difference can be used in the inverse to discover the inputfrequency. The phase contributed error of the FFT to the time-domainoutput signal can thereafter be subtracted out.

A first FFT phase error correction circuit 400 inputs a detecteddifference frequency 402, e.g., from one of the mixers 130 or 134 (FIG.1). An anti-aliasing filter 404 is a low pass filter that cuts off thehighest frequencies to be converted by an ADC 406. A 1024-point FFTfilter 408 is paralleled with a 512-point FFT filter 410. This is onlyan example of the different FFT filters that can be paralleled. Theobjective is to parallel two different FFT filters that will exhibit adifference in their phase responses over the range of frequencies passedby anti-aliasing filter 404. A phase-slope comparator 412 reads thedifferences in the outs of FFT's 408 and 410, and assumes that bothprocessed the same frequency input. So any difference in the FFT outputsis attributed to the respective differences in phase error. Such can beused in a calculation to discern what must be the input frequency toboth FFT's. An FFT phase-frequency correction unit 414 corrects theoutput of the 1024-point FFT filter 408 according to theprocessing-frequency determined by the phase-slope comparator 412. Acorrected time-domain radar output 416 can then be used in conventionalground penetrating radar imaging tomography to visually display objectsburied in the ground.

Referring now to FIG. 5, the FFT contributed phase error can also becorrected with a less processor intensive method than that of FIG. 4.The results obtained with that shown in FIG. 5 are quite good, given theresources demanded. The amplitude responses of adjacent FFT filters canbe compared to discover the input frequency they have both processed. Alookup table is constructed that relates the amplitude difference to thecenter frequency of each adjacent FFT filter. FIG. 3 shows the twodifferent amplitude responses, A and B, of adjacent FFT filters. A lineis drawn between the peaks of each, and the slope of this line willreveal the input frequency from a lookup table.

A second FFT phase error correction circuit 500 inputs a detecteddifference frequency 502, e.g., from one of the mixers 130 or 134 (FIG.1).

An anti-aliasing filter 504 is a low pass filter that cuts off thehighest frequencies to be converted by an ADC 506. A 1024-point FFTfilter 508 is used for amplitude measurements, it will exhibit adifference in its phase responses over the range of frequencies passedby anti-aliasing filter 504. A phase-slope comparator 512 reads thedifferences in the outputs of FFT 508. Any difference in the FFT'soutputs is attributed to the respective differences in phase error. Suchcan be used in a calculation to discern what must be the inputfrequency. An FFT phase-frequency correction unit 514 corrects theoutput of the 1024-point FFT filter 508 according to theprocessing-frequency determined by the phase-slope comparator 512. Acorrected time-domain radar output 516 can then be used in conventionalground penetrating radar imaging tomography to visually display objectsburied in the ground.

In a linear-FM system, every analog-to-digital sample taken over thewhole sweep contributes to the time-domain output signals. In impulseradar systems, each analog-to-digital sample contributes to only onetime-domain output signal sample. If five-hundred analog-to-digitalsamples were taken, the dynamic range of the linear-FM system would bebetter by V500, or about 27-dB. The improved dynamic range provides fora more precise background subtraction so weak target information signalscan be discerned in the presence of strong interfering signals. Sharperimages can be realized with less signal processing because digitalcorrections can be made directly to the radar echo phase and amplitudedata output by the analog-to-digital converter. Otherwise, an inverseFourier transform would have to be made before such corrections could beapplied. Such corrections compensate for system imperfections andprovide higher range resolution and sharper radar images without havingto digitally transform time-domain data into the frequency domain.

DLFM transmitted signals have a high degree repeatability because theyare digitally generated. Such repeatability permits better rejection ofstationary internal clutter signals and an improved visibility ofunderground targets. The amplitude of each waveform sample is digitallycomputed first, then digital-to-analog converted into an analogwaveform. The waveform repeatability is thus that of thedigital-to-analog converter. Demonstrations comparing one radar sweepagainst another sweep show the differences to be only one part in 3000,or about minus 70-dB. Such repeatability enables stationary internalclutter signal cancellation of about 70-dB.

Digital linear-FM radar embodiments of the present invention showreduced radio frequency interference to and from other systems sharingthe same radio frequency spectrum. A typical million-to-one pulsecompression ratio of DLFM radar reduces its susceptibility tointerference from outside in-band signals by 60-dB. Such also reducesthe generated interference in other electronic systems sharing the samespectrum by a similar amount. The DLFM narrowband radar receiver sweepsacross a wide RF spectrum in a short time. An external, spurious signalwould have to sweep at exactly the right rate to interfere with the DLFMreceiver processing. Conversely, the frequency-sweeping, low-powertransmitted signal of the DLFM transmitter delivers so very littleenergy into the passband of another device's receiver that little realsignal power is seen by that receiver. It therefore will not typicallybe disturbed by the operation of a DLFM unit nearby.

Digital linear-FM radar implementations are more reliable because highpeak power transmitted signals are not used. The generation of high peakpower transmitted signals require high operating voltages. Such highvoltages lead to equipment breakdowns and higher construction costs. TheDLFM radar transmitters can operate with 12-volt car battery voltages,or even less voltage.

Digital linear-FM radar units are less susceptible to antenna ringingeffects. The radiated time of each spectral component of theDLFM-waveform is relatively long compared to antenna ringing times,e.g., microseconds versus nanoseconds. So wideband antenna types can beused which would otherwise be inappropriate for use in impulse radarsystems.

Digital linear-FM radars can be programmed to output many differenttypes of waveforms. Different waveforms are useful in gatheringspecialized image and map data, and such help accommodate a wide rangeof terrain and soil conditions. Digital linear-FM radar embodiments ofthe present invention allow the operating program to be adjusted totrade-off image resolution for greater image depth when necessary. Thepower output can also be scaled to respond to soil characteristics.Impulse radars typically radiate only one type of waveform, and thuspreclude a pre-survey adjustment of the probe signal for specific typesof report signals.

Although particular embodiments of the present invention have beendescribed and illustrated, such is not intended to limit the invention.Modifications and changes will no doubt become apparent to those skilledin the art, and it is intended that the invention only be limited by thescope of the appended claims.

1. A radar system, comprising: an antenna array disc (202) with anexternal face and and providing for revolutions in a plane of rotation;a radar antenna (116,118) mounted flat to said external antenna face andfor directing linear frequency-modulated radar transmissions (120)normal to said plane of rotation; a radar processor (140,216,400,500)mounted on said antenna array disc and connected to the radar antenna,and providing for a calculation of electronic range measurements (148)of radar-imaged target objects (122) normal to said radar antenna; and awireless receiver (218) for receiving said range measurements fortomographic imaging of said target objects (122).
 2. The radar system ofclaim 1, further comprising: a motor (212) to rotate the antenna arraydisc; and an angle encoder (214) for determining the instantaneouspositions visited by the radar antenna.
 3. The radar system of claim 1,further comprising: a bowtie element included in the radar antenna(206,208).
 4. The radar system of claim 1, further comprising: a set oforthogonally oriented pairs of vertical and horizontal bowtie elementsall lying in the same plane and included in the radar antenna.
 5. Theradar system of claim 1, further comprising: a switch matrix (112) forselecting amongst a set of orthogonally oriented pairs of vertical andhorizontal bowtie elements all lying in the same plane included with theradar antenna.
 6. The radar system of claim 1, wherein: the radarsignals radiated by the linear-FM radar are narrowband waveforms whichlinearly chirp over a given frequency range rather than radiating allfrequencies at the same instant of time.
 7. The radar system of claim 1,wherein: is provided a reduction in the required analog-to-digitalconversion rate by a factor of approximately one thousand, relative tothat required by an impulse radar.
 8. A digital linearfrequency-modulated radar system, comprising: a linearfrequency-modulated transmitter to transmit a radar signal from a firstantenna to a target; a receiver to receive any echo signal of said radarsignal returned from said target; a mixer to produce a beat tone fromthe instantaneous difference in frequencies of said radar signal andsaid echo signal; an analog-to-digital converter (ADC) to digitallysample said beat tone; a fast Fourier transform (FFT) filter to processfrequency-domain measurements from the ADC into time-domain radar data;and a corrector connected to the FFT and providing for a removal ofphase errors from said time-domain radar data contributed by the FFTfilter that are a function of the frequency of said beat tone.
 9. Thesystem of claim 8, further comprising: an m-point FFT filter to receivedigital samples from the ADC; an n-point FFT filter to receive digitalsamples from the ADC in parallel with the (n) m-point FFT filter; acomparator connected to an output of each of the m-point and n-point FFTfilters and providing for a frequency estimate to said beat toneaccording to a difference in any FFT-output signals produced; and acompensator to receive said frequency estimate and providing for acorrection to said time-domain radar data.
 10. The system of claim 9wherein a plurality of m-point FFT filters receive digital samples fromthe ADC.
 11. The system of claim 9 wherein a plurality of n-point FFTfilters receive digital samples from the ADC.
 12. The system of claim 8,further comprising: a first FFT filter to receive digital samples fromthe ADC; a second FFT filter to receive digital samples from the ADC andadjacent with the first FFT filter a comparator connected to an outputof each of the first and second FFT filters and providing for acorrection to said time-domain radar data based on a difference in thepeak output amplitudes of each.
 13. The system of claim 12, wherein: thecomparator includes a lookup table that provides a correction factor forsaid time-domain radar data based on said difference in the peak outputamplitudes of the first and second FFT filters.
 14. A radar system,comprising: an antenna support disc (202) with an external face forrevolving in a plane of rotation (114); at least one radar antenna,(116,118) mounted flat to said external face for directing linearfrequency-modulated radar transmissions (120) normal to said plane ofrotation; a radar transmitter (102,104,106,108,110) providing linearfrequency-modulated radar transmissions, and connected to such radarantenna (116,118); at least one radar antenna, (126,128) mounted flat tosaid external face for receiving radar transmissions (124) normal tosaid plane of rotation; a radar receiver (130,132,134,136,138) connectedto the radar antenna (126,128); a radar processor (140,216,400,500)connected to the radar receiver, and providing for a calculation ofradar echo range measurements (148) of target objects (122,220,222)illuminated by the radar antenna; and a local computer connected (218)to the radar processor and receiving said range measurements for imagingof said target objects (122,222).
 15. The radar system of claim 14,further comprising: a motor (212) to rotate the antenna array disc; andan angle encoder (214) for determining the instantaneous positionsvisited by the radar antenna.
 16. The radar system of claim 14, furthercomprising: a bowtie element included in the radar antenna (206,208).17. The radar system of claim 14, further comprising: a set oforthogonally oriented pairs of vertically and horizontal bowtie elementsall lying in the same plane and included in the radar antenna.
 18. Theradar system of claim 14, further comprising: a switch matrix (112) forselecting amongst a set of orthogonally oriented pairs of vertical andhorizontal bowtie elements all lying in the same plane included with theradar antenna.
 19. The radar system of claim 14 wherein the radarantenna, is mounted flat to the external face and directs non-linearfrequency-modulated radar transmissions normal to the plane of rotation.